Despite their lack in resolution and accuracy in comparison to digital readings, analogue moving-oil meters remain the display of choice when it comes to tracking a reading’s trend or drawing information upon a measurements rate of change.
For low-level current measurements, however, the meter current for a full-scale deflection usually exceeds the current to be measured, and a separate supply driving the meter is required. Analogue meters of the past, such as the Multavi-10 from Hartmann & Braun, solved this problem by implementing a rechargeable accumulator as the meter supply. Manually selectable shunt resistors in combination with a high-precision chopper amplifier allowed the user to choose from 13 different current ranges between 1µA and 1A.
With the introduction of modern current-shunt monitor ICs, such as the INA19x family, the amplifier design of moving-coil meters has been drastically simplified. Fig. 1 shows the drive circuit of an 8-in moving-coil meter, measuring a current range from 0 to 100mA. The meter current for a full-scale deflection is 15mA. The current-shunt monitor, INA193, senses the voltage drop across the 1Ω shunt resistor, RS1. At a maximum current of 100mA the voltage across RS1 is 100mV.
The value chosen for RS1 depends on the application and is a compromise between small-signal accuracy and maximum permissible voltage drop in the measurement line. High values of RS1 provide better accuracy at lower currents by minimising the effects of offset, while low values of RS1 minimise voltage loss in the supply line. For most applications, best performance is attained with an RS1 value that provides a full-scale shunt voltage range of 50mV to 100mV. The maximum input voltage for accurate measurements is 500mV.
In this example, the INA193 amplifies the 100mV full-scale input by a gain-factor of 20V/V, thus providing a full-scale output of 2V. The succeeding operational amplifier, OPA344, possesses rail-to-rail inputs and outputs, and operates in conjunction with the N-channel MOSFET, BSN254, as a voltage-controlled current-source.
The entire meter circuit, including the INA193, operates from a single 5V supply, which also limits the maximum output voltage swing of the OPA to 5V. It is necessary to choose a MOSFET with a low Gate-Source threshold voltage, VGS, since this voltage subtracts from the OPA output swing. The BSN254 has a maximum threshold voltage of 2V, which suffices the low-VGS requirement. Because the voltage at the non-inverting OPA input equals the one at the inverting input, the
full-scale output of 2V lies across RS2. To allow the maximum deflection current to flow, RS2 is calculated via:
Adjust RS2 to calibrate the meter or to change its full-scale current range. Adjust RS1 to increase low-current measurement accuracy or to extend the measurement range to higher current values. Another benefit of the circuit is, that you can separate the meter from the point of measurement. Moving-coil meters are not intended for high-precision measurements, so relaxed-accuracy resistors can be used. Bypassing the instrument supply with decoupling capacitors is necessary to avoid stray pick-up from the electrical-noise environment.
The INA193 is just one member of a family of current-shunt monitors. The INA194 and INA195 are members with the same pin-out, but provide different gains of 50V/V and 100V/V respectively. A further trio of current-shunt monitors, the INA196, INA197 and INA198 are functionally identical, but come in a different pin-out.
The INA19x family uses a new internal circuit topology that provides common-mode range extending from -16V to +80V while operating from a single power supply. The common-mode rejection in a classic instrumentation amp approach is limited by the requirement for accurate resistor matching. By converting the induced input voltage to a current, the INA19x provides common-mode rejection that is no longer a function of closely matched resistor values, providing the enhanced performance necessary for such a wide common-mode range.
The simplified diagram in Fig.2 shows the basic circuit function. When the common-mode voltage is positive, amplifier A2 is active. The differential input voltage, (Vin+) – (Vin-) applied across RS, creates the voltage potentials, vN and vP at A2’s inputs:
vn = Vin+ -Is . Rs and vp = Vin+
To make vp = vn, A2 must drive the transistor so, that its collector current, IC, causes a voltage drop across the 5kΩ resistor, that equals the differential input voltage:
vp = vn
Vin+ -Ic . 5k = Vin+ -Is . Rs
Ic . 5k = Is . Rs
Expressing IC through the ratio of output voltage to load resistor, IC = Vout/Rl, defines output voltage as:
Vout = Is . Rs·Rl/5k.
When the common-mode voltage is negative, amplifier A1 is active. The differential input voltage, (Vin+) – (Vin .) applied across Rs, is converted to a current through a 5kΩ resistor. This current is sourced from a precision current mirror whose output is directed into RL converting the signal back into a voltage and amplified by the output buffer amplifier. Circuit architecture ensures smooth operation, even during transition where both amplifiers A1 and A2 are active.
The input pins, Vin+ and Vin-, should be connected closely to the shunt resistor to minimise any resistance in series with the shunt resistance. Power-supply bypass capacitors are required for stability. Applications with noisy or high impedance power supplies may require additional decoupling capacitors to reject power-supply noise. Connect bypass capacitors close to the device pins.
The input circuitry of the INA19x can accurately measure beyond its power-supply voltage, V+. For example, the V+ power supply can be 5V, whereas the load power-supply voltage is up to +80V. The output voltage range of the OUT terminal, however, is limited by the voltages on the power-supply pin.
The output of the INA19x is accurate within the output voltage swing range set by the power supply pin, V+. This is best illustrated when using the INA195 or INA198 (which are both versions using a gain of 100), where a 100mV
full-scale input from the shunt resistor requires an output voltage swing of +10V, and a power-supply voltage sufficient to achieve +10V on the output.
An obvious and straightforward location for filtering is at the output of the INA19x series; however, this location negates the advantage of the low output impedance of the internal buffer. The only other option for filtering is at the input pins of the INA19x, which is complicated by the internal 5kΩ + 30percent input impedance; see Fig.2. Using the lowest possible resistor values minimises both the initial shift in gain and effects of tolerance. The effect on initial gain is given by:
Total effect on gain error can be calculated by replacing the 5k term with 5kΩ -30percent , (or 3.5kΩ) or 5kΩ +30percent (or 6.5kΩ). The tolerance extremes of RFILT can also be inserted into the equation. If a pair of 100Ω 1percent resistors are used on the inputs, the initial gain error will be 1.96percent.
Worst-case tolerance conditions will always occur at the lower excursion of the internal 5kΩ resistor (3.5kΩ), and the higher excursion of Rfilt -3percent in this case.
Thomas Kugelstadt is with Texas Instruments Ltd, Northampton, UK. www.ti.com"